Cellular CDMA notch filter

ABSTRACT

A notch filtering apparatus comprises an input configured to receive at least one cellular code division multiple access (CDMA) signal. Each received cellular CDMA signal has an associated spread bandwidth. A notch filter device is configured to attenuate a bandwidth at a plurality of frequencies within the associated spread bandwidth of the at least one received cellular CDMA signal.

This application is a continuation of application Ser. No. 09/878,586,filed Jun. 11, 2001, which is a continuation of application Ser. No.09/400,715, filed Sep. 21, 1999, which issued May 14, 2002 as U.S. Pat.No. 6,389,002, which is a continuation of application Ser. No.08/951,058, filed Oct. 15, 1997, which issued on Jan. 4, 2000 as U.S.Pat. No. 6,011,789, which is a continuation of application Ser. No.08/329,371, filed Apr. 7, 1994, which issued on Dec. 30, 1997 as U.S.Pat. No. 5,703,874, which is a continuation of application Ser. No.07/983,070, filed Nov. 25, 1992, now abandoned, which is acontinuation-in-part of application Ser. No. 07/622,235, filed Dec. 5,1990, which issued on Sep. 27, 1994 as U.S. Pat. No. 5,351,269.

BACKGROUND

This invention relates to spread-spectrum communications and moreparticularly to a broadband code division multiple access communicationssystem which communicates over the same frequency band of an existingfrequency division multiple access (FDMA), proposed time divisionmultiple access (TDMA) or any other mobile-cellular system.

DESCRIPTION OF THE RELEVANT ART

The current mobile-cellular system uses the frequency band 868-894 MHzfor transmission from the mobile unit to the cellular base stations andthe frequency band 823-849 MHz for transmission from the cellular basestations to the mobile unit. Each of these frequency bands is divided inhalf to permit two competitive systems to operate simultaneously. Thus,each system has 10.0 MHz available for transmission and 10.0 MHz forreception. Each of the 10.0 MHz bands is divided into 30 kHz channelsfor voice communications.

A problem in the prior art is limited capacity due to the number ofchannels available in the mobile radio cellular system.

FIG. 1 is a diagram of the cellular system. A mobile unit serviced bycell A located near the border of cells A and B and a mobile unitserviced by cell B located near the same border are received by thecellular base stations of cells A and B with almost the same power. Toavoid interference between units operating in the same frequency band atcomparable power levels, different frequency subbands, i.e. channels,are allocated to adjacent cells. FIG. 1 shows a seven frequency scheme,with each cell having a bandwidth equaling 10.0 MHz/7, whichapproximately equals 1.4 MHz. This frequency scheme has adjacent cellsoperating at different frequencies, thereby reducing interference amongunits in adjacent cells. This technique is called frequency reuse. As aresult of frequency reuse, each cell has N=1.4 MHz/30 kHz=46 channels,with the channels divided and allocated as shown in FIG. 2. The 1.4 MHzis divided for FDMA or TDMA into 30 kHz channels for communications,each with a 180 kHz guard band. Each of the different cells, A throughG, have channels that lie in a different 30 kHz band so that theirspectra do not overlap. FIG. 2 also shows the spread-spectrum signaloverlaying on the existing units whether they be Advanced Mobile PhoneService (AMPS) or IS54. Some of these channels are reserved forsignaling, leaving approximately 41 channels per cell. The channels areallocated to cells A, B, and C as shown in FIG. 2. A guard band of 180kHz separates each channel so that adjacent channel units within thesame cell do not interfere with one another.

The cells in a mobile-cellular system are expensive to maintain, andprofitability can be significantly increased by increasing the number ofunits per cell. One approach to increase the number of units per cell isto change from analog frequency modulation (FM) communication, and touse digital communication with time division multiple access.

Existing AMPS cellular systems exhibit numerous deficiencies; forexample, lack of privacy, dropped calls, low data rate capabilities,.poor quality speech, and limited capacity. TDMA and code divisionmultiple access (CDMA) proposals attempt to overcome the capacity andprivacy issues, with TDMA systems proposing to pack either 3 or 6 unitsin each 30 kHz frequency band previously used by the AMPS FM unit, whilenarrowband CDMA proposals, for example, by Qualcomm, claim a furtherincreased capacity.

SUMMARY

A notch filtering apparatus comprises an input configured to receive atleast one cellular code division multiple access (CDMA) signal. Eachreceived cellular CDMA signal has an associated spread bandwidth. Anotch filter device is configured to attenuate a bandwidth at aplurality of frequencies within the associated spread bandwidth of theat least one received cellular CDMA signal.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated in and constitute apart of the specification, illustrate preferred embodiments of theinvention, and together with the description serve to explain theprinciples of the invention.

FIG. 1 illustrates a seven-frequency-set mobile-cellular plan;

FIG. 2 shows cellular channels which are separated by a guard band of180 kHz;

FIG. 3 is a block diagram of a spread-spectrum-base station receiver;

FIG. 4 is a block diagram of a first embodiment of aspread-spectrum-base station transmitter;

FIG. 5 is a block diagram of a second embodiment of aspread-spectrum-base station transmitter;

FIG. 6 is a block diagram of a unit-spread-spectrum receiver;

FIG. 7 is a block diagram of a first embodiment of unit-spread-spectrumtransmitter;

FIG. 8 is a block diagram of a second embodiment of aunit-spread-spectrum transmitter;

FIG. 9 shows the spectrum of a spread-spectrum signal with an AM signalof equal power at its carrier frequency;

FIG. 10 shows a spread-spectrum data signal when the spread-spectrumsignal power is equal to an AM signal power;

FIG. 11 shows an audio signal when the spread-spectrum signal power isequal to the AM signal power;

FIG. 12 shows a pseudo-random sequence generator;

FIG. 13 shows position settings of switches of FIG. 12 to form PNsequences;

FIG. 14 illustrates a phasor diagram for the phase of a CDMA interferer;

FIG. 15 plots output signal-to-noise ratios for different numbers ofCDMA units in an AMPS antenna sector;

FIG. 16 illustrates a coherent demodulator for performing a demodulationprocedure;

FIG. 17 shows rectangular pulses c_(o) and c′_(i);

FIG. 18 shows a half-sine wave pulse c′_(i)(t)=sin [πt/(Tc)] and arectangular pulse c_(o)(t);

FIG. 19 illustrates the effect of AMPS units on a B-CDMA base station;

FIG. 20 illustrates the effect of B-CDMA units on an AMPS base station;

FIG. 21 illustrates the effect of B-CDMA base stations on CDMA units;

FIG. 22 illustrates the effect of AMPS base stations on CDMA units;

FIG. 23 shows the effect of the CDMA base stations on AMPS units;

FIG. 24 illustrates the use of a comb filter;

FIG. 25 shows a cellular system employing 6-segment antennas; and

FIG. 26 illustrates a region having a notching-out of power.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT(S)

Reference is now made in detail to the present preferred embodiments ofthe invention, examples of which are illustrated in the accompanyingdrawings, wherein like reference numerals indicate like elementsthroughout the several views.

The spread-spectrum code division multiple access (CDMA) communicationssystem of the present invention is located within a same geographicalregion, i.e. cell, as occupied by a mobile-cellular system. Each cell ofthe mobile-cellular system has a cellular bandwidth. In presentlydeployed mobile-cellular systems, the cellular bandwidth isapproximately 10.0 MHz. The cellular bandwidth is divided into aplurality of predetermined channels. Each predetermined channeltypically has a bandwidth of 30 kHz. The predetermined channels areseparated by guard bands. The usual guard band separation is 180 kHz.Cellular units communicate on the predetermined channels, currentlyusing frequency modulation (FM).

The spread-spectrum CDMA communications system includes a plurality ofspread-spectrum-base stations and a plurality of spread-spectrum unitslocated within the same geographical region, i.e. cell, as occupied bythe mobile-cellular system. The spread-spectrum CDMA communicationssystem can be used for communicating data between a plurality ofspread-spectrum units and the spread-spectrum-base station. The data maybe, but are not limited to, computer data, facsimile data or digitizedvoice.

A spread-spectrum-base station, which preferably is collocatedgeographically with a cellular-base station, communicates data betweenthe plurality of spread-spectrum units, with a first spread-spectrumuser uses a first spread-spectrum unit, and a second spread-spectrumuser uses a second spread-spectrum unit, etc.

Each spread-spectrum-base station includes at least one set ofbase-converting means, base-spread-spectrum-processing means,base-transmitting means, base-detection means and a base antenna. Eachspread-spectrum base station additionally includes base-comb-filtermeans and/or base-sector means. The term “base” is used as a prefix toindicate that a respective element is located at thespread-spectrum-base station.

The base-comb-filter means may include notch filters which attenuate themobile-cellular signal power transmitted on predetermined channels ofthe mobile-cellular system. The base-detection means may includebase-despreading means and base-synchronizing means. The base-detectionmeans broadly converts data communicated from a spread-spectrum unitinto a form for output to a user. The base-despreading means broadlydespreads a received spread-spectrum signal.

The base-comb-filter means, as shown in FIG. 3, may be embodied as acomb filter 140. The comb filter 140 notches the predetermined channelsof the mobile-cellular system. The comb filter 140 reduces the combinedinterfering power level from mobile-cellular units with thespread-spectrum-base station. For the presently deployed mobile-cellularsystem, by way of example, the comb filter 140 serves as a plurality ofnotch filters, blocking the 30 kHz bandwidth at each frequency locationof the predetermined channels of the mobile-cellular system.

The base-despreading means, as illustrated in FIG. 3, may be embodied asa pseudorandom generator, a plurality of product devices 141 and aplurality of bandpass filters 143. The pseudorandom generator storeschip codes, g₁(t), g₂(t), . . , g_(N)(t), for demodulating data fromspread-spectrum signals received from the plurality of spread-spectrumunits at the spread-spectrum-base station. The base-detection means alsoincludes base-synchronizing means for synchronizing the base-despreadingmeans to received spread-spectrum signals. The base-synchronizing meansmay be embodied as a phase-locked-loop circuit.

Alternatively, the base-despreading means may be embodied as a pluralityof matched filters. The plurality of matched filters may employsurface-acoustic-wave (SAW) devices, digital signal processors, or acombination of analog and digital technologies, as is well known in theart. Each matched filter has an impulse response matched to each chipcode g₁(t), g₂(t), . . . , g_(N)(t), respectively, for demodulating datafrom the spread-spectrum signals received from the plurality ofspread-spectrum units. The impulse responses of the matched filters maybe either fixed or programmable.

The spread-spectrum receiver at the spread-spectrum-base stationprocesses selected data received from a selected spread-spectrum unit,which were transmitted with a spread-spectrum signal using aselected-chip-code, g_(i)(t). The detector 145 demodulates the selecteddata from the despread spread-spectrum signal.

A plurality of product devices 141, bandpass filters 143 and detectors145 may be coupled through a power splitter 147 to an antenna 149, forreceiving simultaneously multiple spread-spectrum channels. Each productdevice 141 uses a selected chip code for demodulating a selectedspread-spectrum signal, respectively.

For a spread-spectrum system to operate properly, the spread-spectrumreceiver must acquire the correct phase position of the received spreadspectral signal, and the spread-spectrum receiver must continually trackthat phase position so that loss-of-lock does not occur. The twoprocesses of acquisition and tracking form the synchronization subsystemof the spread-spectrum receiver. The former operation typically isaccomplished by a search of as many phase positions as necessary untilone phase position is found which results in a large correlation betweenthe phase of the incoming signal and the phase of the locally generatedspreading sequence at the receiver. This former process typically occursusing a correlator or a matched filter. The latter operation is oftenperformed with a delay-locked loop. The importance of the combinedsynchronization process can not be over stated for if synchronization isnot both achieved and maintained, the desired signal cannot be despread.

The base-converting means, as illustrated in FIG. 4, may be embodied asa base modulator 151. Each base modulator 151 converts the format ofdata to be transmitted to a spread-spectrum unit into a form forcommunicating over radio waves. For example, an analog voice signal maybe converted to a base-data signal, using a technique called sourceencoding. Typical source coders are linear predictive coders, vocoders,delta modulators and pulse code modulation coders.

The base-spread-spectrum-processing means may be embodied as abase-spread-spectrum modulator 153. Each base-spread-spectrum modulator153 is coupled to a respective base modulator 151. Eachbase-spread-spectrum modulator 153 modulates the converted-data signalusing spread-spectrum. The converted data is multiplied using a productdevice, or is modulo-2 added, using an EXCLUSIVE-OR gate 153, with aselected spread-spectrum chip code, g_(N)+i(t).

The base-transmitter means may be embodied as a base transmitter 155.Each base transmitter 155 is coupled to a respectivebase-spread-spectrum modulator 153. Each base transmitter 155 transmits,across the cellular bandwidth, the spread-spectrum-processed-converteddata from the spread-spectrum-base station to a spread-spectrum unit.Each base transmitter 155 also modulates thespread-spectrum-processed-converted data at a carrier frequency, f_(o).

Each base transmitter 155 has a transmitter oscillator which supplies acarrier signal having a carrier frequency. The transmitter oscillator iscoupled to a transmitter-product device. The base transmitter 155multiplies, using the transmitter-product device, thespread-spectrum-processed-converted data by the carrier signal.

The base-transmitting means may, in a preferred embodiment, transmitdata using a spread-spectrum signal having a power level limited to apredetermined level. The base-transmitting means may also transmit databy adding the plurality of spread-spectrum data signals.

A plurality of base modulators 151, base product devices 153 and basetransmitters 155 may be coupled through a power combiner 157 to anantenna 159 for simultaneously transmitting a multiplicity ofspread-spectrum channels. FIG. 4 is an illustrative embodiment forgenerating simultaneous spread-spectrum signals, and there are manyvariants for interconnecting product devices, modulators andtransmitters, for accomplishing the same function.

As an alternative example, FIG. 5 illustrates a spread-spectrum-basestation transmitter which may be used for producing the same result asthe transmitter of FIG. 4. The base-spread-spectrum modulator at thespread-spectrum-base station includes means for processing data forparticular spread-spectrum units with a selected chip code. In FIG. 5,the base-spread-spectrum-processing means may be embodied asEXCLUSIVE-OR gates 253, with the data being modulo-2 added, usingEXCLUSIVE-OR gates 253, with a selected spread-spectrum chip code,g_(N)+i(t). The resulting spread-spectrum processed data from aplurality of EXCLUSIVE-OR gates 253 are combined using combiner 257. Thebase transmitter 255 modulates the combined spread-spectrum-processeddata at the carrier frequency, f_(o). The base transmitter 255 iscoupled to the antenna 159 and simultaneously transmits the plurality ofspread-spectrum-processed data as a spread-spectrum signal having amultiplicity of spread-spectrum channels.

One or more spread-spectrum-base stations may include base-sector means.The base sector means includes the control circuits and base antennasfor dividing a geographical coverage area of the spread-spectrum-basestation into two, three, four, five, six, or more sectors. Thegeographical coverage area, by way of example, may be divided into threesectors of 120°, by using three antennas with appropriate beamwidths.Each antenna is coupled to separate sets of base-converting means,base-spread-spectrum-processing means, base-transmitting means, andbase-detection means. The control circuits include the capability forhanding-off a spread-spectrum unit traversing from a first sector to asecond sector of the spread-spectrum-base station. Circuits forperforming this hand-off are described in U.S. patent application byDonald L. Schilling, entitled SPREAD SPECTRUM CELLULAR HANDOFF APPARATUSAND METHOD, having Ser. No. 07/727,617, filing date of Jul. 10, 1991,now issued U.S. Pat. No. 5,179,571, and incorporated herein byreference. The sector means allows lower power levels to be transmittedfrom the spread-spectrum-base station, by focusing the required powerinto the sector in which the spread-spectrum unit is located. Thissectoring of the geographical area reduces the combined interferingpower level from the mobile-cellular units with the spread-spectrum-basestation.

The present invention also includes spread-spectrum units which arelocated within the cell. Each of the spread-spectrum units has a unitantenna, unit-spread-spectrum-detection means, unit-converting means,unit-spread-spectrum-processing means, and unit-transmitting means. Theunit-detection means is coupled to the unit antenna. The unit-detectionmeans includes unit-despreading means. The term “unit” is used as aprefix to indicate that a respective element is located at aspread-spectrum unit.

The unit-detection means recovers data communicated to thespread-spectrum unit from the spread-spectrum-base station. Theunit-detection means also includes means for converting the format ofthe data into a form suitable for output to a user. The format may be,for example, computer data, an analog speech signal or otherinformation. The unit-detection means, by way of example, may includetracking and acquisition circuits for the spread-spectrum signal, aproduct device for despreading the spread-spectrum signal, and anenvelope detector. FIG. 6 illustratively shows unit-detection meansembodied as a unit-spread-spectrum receiver including aunit-spread-spectrum demodulator 161, a unit-bandpass filter 163, and aunit-data detector 165, coupled to an antenna 169.

The unit-spread-spectrum demodulator 161 despreads using a chip-codesignal having the same or selected chip code, g_(N)+i(t), as thereceived spread-spectrum signal, the spread-spectrum signal receivedfrom the spread-spectrum-base station. The unit-bandpass filter 163filters the despread signal and the unit-data detector 165 puts theformat of the despread spread-spectrum signal into a form suitable foroutput to a user of the spread-spectrum unit.

The unit-despreading means includes means for storing a local chip code,g_(N)+i(t), for comparing the local chip code to signals received forrecovering data sent from the spread-spectrum-base station to thespread-spectrum unit. The storing means may be embodied as a memory.

The unit-detection means also may include unit-synchronizing means forsynchronizing the unit-despreading means to received signals. Theunit-synchronizing means may be embodied as a phase-lock loop circuit.Similarly, the base-despreading means at the spread-spectrum-basestation includes means for processing data for particularspread-spectrum units with a selected chip code. The unit-despreadingmeans may be embodied as EXCLUSIVE-OR gates for modulo-2 adding the datawith a selected spread-spectrum chip code.

Alternatively, the unit-despreading means may be embodied as a matchedfilter. The matched filter may employ a surface-acoustic-wave (SAW)device, a digital signal processor, or a combination of analog anddigital technologies, as is well known in the art. Each matched filterhas an impulse response matched to each chip code g₁(t), g₂(t), . . . ,g_(N)(t), respectively, for demodulating data from the spread-spectrumsignals received from the base station. The impulse responses of thematched filters may be either fixed or programmable.

The unit-converting means, as illustrated in FIG. 7, may be embodied asa unit modulator 171. The unit modulator 171 converts the format of thedata into a form suitable for communicating over radio waves. Similar tothe spread-spectrum-base station, an analog voice signal may beconverted to a converted-data signal, using a technique called sourceencoding. As with the base modulator 151, typical source encoders arelinear predictive coders, vocoders, delta modulators and pulse codemodulation.

The unit-spread-spectrum-processing means may be embodied as aunit-spread-spectrum modulator 173. The unit-spread-spectrum modulator173 is coupled to the unit modulator 171. The unit-spread-spectrummodulator 173 modulates the converted-data signal with a selected chipcode, g_(i)(t). The converted-data signal is multiplied using a productdevice, or is modulo-2 added, using an EXCLUSIVE-OR gate, with theselected chip code, g_(i)(t).

The unit-transmitting means in FIGS. 7 and 8 may be embodied as a unittransmitter 175. The unit transmitter 175 transmits across the cellularbandwidth, the spread-spectrum-processed-converted data from thespread-spectrum unit to the spread-spectrum-base station. The unittransmitter 175 modulates the spread-spectrum-processed-converted dataat a carrier frequency, f_(o). The carrier frequency of the unittransmitter and the base transmitter may be at the same or differentfrequencies.

As an equivalent transmitter, FIG. 8 illustrates a transmitter for aspread-spectrum unit having unit-spread-spectrum-processing means as amodulo-2 adder, embodied as an EXCLUSIVE-OR gate 273. The EXCLUSIVE-ORgate 273 modulo-2 adds the converted data signal with the selected chipcode, g_(i)(t).

Each of the embodiments of the unit-spread-spectrum transmitter in FIGS.7 and 8 may include a unit transmitter 175. The unit transmitter 175 iscoupled between the unit-spread-spectrum modulator 173 and unit antenna179, as in FIG. 7, or, in the alternate embodiment, the unit transmitter175 is coupled between the EXCLUSIVE-OR gate 273 and the unit antenna179. The unit transmitter 175 transmits across the cellular bandwidth,the spread-spectrum-processed-converted data from the spread-spectrumunit to the spread-spectrum-base station.

A key to the present invention is that the spread-spectrum signals aredesigned to be transparent to other units, i.e. spread-spectrum signalsare designed for negligible interference to the communication of other,existing units. The presence of a spread-spectrum signal is difficult todetermine. This characteristic is known as low probability ofinterception (LPI) and low probability of detection (LPD). The LPI andLPD features of spread-spectrum allow transmission between units of aspread-spectrum CDMA communications system without the existing units ofthe mobile-cellular system experiencing significant interference. Thepresent invention makes use of LPI and LPD with respect of thepredetermined channels using FM in a mobile-cellular system. By havingthe power level of each spread-spectrum signal below the predeterminedlevel, the total power from all spread-spectrum used within a cell doesnot interfere with units in the mobile-cellular system.

In addition, spread-spectrum is also jam or interference resistant. Aspread-spectrum receiver spreads the spectrum of the interfering signal.This reduces the interference from the interfering signal so that itdoes not noticeably degrade performance of the spread-spectrum system.This feature of interference reduction makes spread-spectrum useful forcommercial communications, i.e. the spread-spectrum waveforms can beoverlaid on top of existing narrowband signals.

The present invention employs direct sequence spread-spectrum, whichuses a phase modulation technique. Direct sequence spread-spectrum takesthe power that is to be transmitted and spreads it over a very widebandwidth so that the power per unit bandwidth (Watts/Hertz) isminimized. When this is accomplished, the transmitted spread-spectrumpower received by a mobile-cellular unit, having a relatively narrowbandwidth, is only a small fraction of the actual transmitted power.

In a mobile-cellular system, by way of example, if a spread-spectrumsignal having a power of 10 mw is spread over a cellular bandwidth of10.0 MHz and a cellular unit employs a communication system having achannel bandwidth of only 30 kHz, then the effective interfering powerdue to one spread-spectrum signal, in the narrow band communicationsystem, is reduced by the factor of 10.0 MHz/30 kHz which isapproximately 330. Thus, the effective interfering power is 10 mWdivided by 330 or 0.030 mW. For fifty concurrent units ofspread-spectrum, the power of the interfering signal due tospread-spectrum is increased by fifty to a peak interfering power of1.00 mW.

The feature of spread-spectrum that results in interference reduction isthat the spread-spectrum receiver actually spreads the received energyof any interferer over the same wide bandwidth, 10.0 MHz in the presentexample, while compressing the bandwidth of the desired received signalto its original bandwidth. For example, if the original bandwidth of thedesired spread-spectrum data signal is only 30 kHz, then the power ofthe interfering signal produced by the cellular base station is reducedby 10.0 MHz/30 kHz which is approximately 330.

Direct sequence spread-spectrum achieves a spreading of the spectrum bymodulating the original signal with a very wideband signal relative tothe data bandwidth. This wideband signal is chosen to have two possibleamplitudes, +1 and −1, and these amplitudes are switched, in apseudo-random manner, periodically. Thus, at each equally spaced timeinterval, a decision is made as to whether the wideband modulatingsignal should be +1 or −1. If a coin were tossed to make such adecision, the resulting sequence is truly random. However, in such acase, the receiver is does not know the sequence beforehand and couldnot properly receive the transmission. Instead, a chip-code generatorgenerates electronically an approximately random sequence, called apseudo-random sequence, which is known beforehand to both thetransmitter and the receiver.

To illustrate the characteristics of spread-spectrum, consider 4800 bpsdata which are binary phase-shift keyed (BPSK) modulated. The resultingsignal bandwidth is approximately 9.6 kHz. This bandwidth is then spreadusing direct sequence spread-spectrum to 16 MHz. Thus, the processinggain, N, is approximately 1600 or 32 dB.

Alternatively, consider a more typical implementation with 4800 bps datawhich is modulo-2 added to a spread-spectrum-chip-code signal, g_(i)(t),having a chip rate of 8 Mchips/sec. The resulting spread-spectrum dataare binary phase-shift keyed (BPSK) modulated. The resultingspread-spectrum bandwidth is 16 MHz. Thus, the processing gain is:N=(8×10⁶)/(4.8×10³), which approximately equals 1600, or 32 dB.

FIG. 9 shows the spectrum of this spread-spectrum signal on an amplitudemodulated 3 kHz sinusoidal signal, when they each have the same powerlevel. The bandwidth of the AM waveform is 6 kHz. Both waveforms havethe same carrier frequency.

FIG. 10 shows the demodulated square-wave data stream. This waveform hasbeen processed by an integrator in the receiver, hence the triangularshaped waveform. Note that positive and negative peak voltagesrepresenting a 1-bit and 0-bit are clearly shown. FIG. 11 shows that thedemodulated AM signal replicates the 3 kHz sine wave.

The AM signal does not degrade the reception of data because thespread-spectrum receiver spreads the energy of the AM signal over 16MHz, while compressing the spread-spectrum signal back to its original9.6 kHz bandwidth. The amount of the spread AM energy in the 9.6 kHzBPSK bandwidth is the original energy divided by N=1600; or,equivalently, it is reduced by 32 dB. Since both waveforms initiallywere of equal power, the signal-to-noise ratio is now 32 dB, which issufficient to obtain a very low error rate.

The spread-spectrum signal does not interfere with the AM waveformbecause the spread-spectrum power in the bandwidth of the AM signal isthe original power in the spread-spectrum signal divided by N₁, whereN₁=16 MHz/6 kHz=2670, or 33 dB; hence, the signal-to-interference ratioof the demodulated sine wave is 33 dB.

The direct sequence modes of spread-spectrum uses pseudo randomsequences to generate the spreading sequence. While there are manydifferent possible sequences, the most commonly used are maximal-lengthlinear shift register sequences, often referred to as pseudo noise (PN)sequences. FIG. 12 shows a typical shift register sequence generator.FIG. 13 indicates the position of each switch b_(i) to form a PNsequence of length L, whereL=2^(N)−1

The characteristics of these sequences are indeed noise like. To seethis, if the spreading sequence is properly designed, the spreadingsequence has many of the randomness properties of a fair coin tossexperiment where 1=heads and −1=tails. These properties include thefollowing:

-   -   1) In a long sequence, about ½ of the chips are +1 and ½ of the        chips are −1.    -   2) The length of a run of r chips of the same sign occurs about        L/2^(r) times in a sequence of L chips.    -   3) The autocorrelation of the sequence PN_(i)(t) and PN_(i)(t+τ)        is very small except in the vicinity of =0.    -   4) The cross-correlation of any two sequences PN_(i)(t) and        PN_(j)(t+τ) is small.

Code Division Multiple Access

Code division multiple access is a direct sequence spread-spectrumsystem in which a number, at least two, of spread-spectrum signalscommunicate simultaneously, each operating over the same frequency band.In a CDMA system, each unit is given a distinct chip code. This chipcode identifies the unit. For example, if a first unit has a first chipcode, g₁(t), and a second unit a second chip code, g₂(t), etc., then areceiver, desiring to listen to the first unit, receives at thereceiver's antenna all of the energy sent by all of the units. However,after despreading the first unit's signal, the receiver outputs all theenergy of the first unit but only a small fraction of the energies sentby the second, third, etc., units.

CDMA is interference limited. That is, the number of units that can usethe same spectrum and still have acceptable performance is determined bythe total interference power that all of the units, taken as a whole,generate in the receiver. Unless one takes great care in power control,those CDMA transmitters which are close to the receiver cause theoverwhelming interference. This effect is known as the near-far problem.In a mobile environment the near-far problem could be the dominanteffect. Controlling the power of each individual mobile unit is possibleso that the received power from each mobile unit is the same. Thistechnique is called adaptive power control. See U.S. Pat. No. 5,093,840,issued Mar. 3, 1992, entitled, ADAPTIVE POWER CONTROL FOR ASPREAD-SPECTRUM TRANSMITTER, by Donald L. Schilling, which isincorporated herein by reference.

It has been proposed to set aside 10% of the mobile-cellular bandwidth,or 1.00 MHz, to employ CDMA, for eliminating 10% of the currentlyexisting mobile-cellular channels, i.e. approximately 5 channels,thereby restricting the use and access of present subscribers to themobile-cellular system. Further, such a procedure disrupts currentservice as the base station of each cell must be modified.

As a result of this procedure, the existing units are penalized, sincethe number of available channels are reduced by 10% and a cellularcompany employing this approach must modify each cell by firsteliminating those channels from use and then installing the new CDMAequipment.

The present invention is for a CDMA system which does not affectexisting units in so far as it does not require that 10% of the band beset aside. Indeed, using this invention, an entirely separate CDMAsystem can be inserted into the existing mobile spectrum withoutaffecting the existing operation of the frequency division multipleaccess (FDMA) mobile-cellular system or the forthcoming time divisionmultiple access (TDMA) system.

The Proposed Spread-Spectrum CDMA System

The spread-spectrum communications system of the present invention is abroadband CDMA system. Spread-spectrum CDMA can significantly increasethe number of units per cell, compared to TDMA. With CDMA, each unit ina cell uses the same frequency band. However, each spread-spectrum CDMAsignal has a separate pseudo random code which enables a receiver todistinguish a desired signal from the remaining signals. Spread-spectrumunits in adjacent cells use the same frequency band and the samebandwidth, and therefore interfere with one another. A received signalmay appear somewhat noisier as the number of units' signals received bya spread-spectrum base station increases.

Each unwanted unit's signal generates some interfering power whosemagnitude depends on the processing gain. Spread-spectrum units inadjacent cells increase the expected interfering energy compared tospread-spectrum units within a particular cell by about 50%, assumingthat the spread-spectrum units are uniformly distributed throughout theadjacent cells. Since the interference increase factor is not severe,frequency reuse is not employed. Each spread-spectrum cell can use afull 10.0 MHz band for transmission and a full 10.0 MHz band forreception. Hence, using a chip rate of six million chips per second anda coding data rate of 4800 bps results in approximately a processinggain of 1250 chips per bit. It is well known to those skilled in the artthat the number of spread-spectrum units is approximately equal to theprocessing gain. Thus, up to 1250 units can operate in the 10.0 MHzbandwidth of the mobile-cellular system.

The following shows that the B-CDMA spread-spectrum system, with a chiprate of 10 Mchips/sec and with the 3-CDMA spread-spectrum systemoccupying the full cellular band, can overlay on the existing AMPSsystem, and allows an additional 496 B-CDMA spread-spectrum units to the50 AMPS mobile-cellular units located in the 10 MHz band. If TDMA wereemployed rather than AMPS, then 496 B-CDMA spread-spectrum units couldequally overlay the 150 TDMA units. The calculations presented belowassume that the AMPS system uses a three-sector antenna system. Asix-sector system increases capacity. In addition, in the preferredembodiment, the B-CDMA spread-spectrum system uses a six-sector antennasystem, as shown in FIG. 25, since using three-sectors decreasescapacity.

Frequency Modulation

Consider the AMPS mobile-cellular unit receiving the desired FM signalplus interfering B-CDMA spread-spectrum signals. The input to thediscriminator is therefore:${v_{i}(t)} = {{\sqrt{2P_{A}}{\cos\left( {{\omega_{0}t} + {\theta(t)}} \right)}} + {\sum\limits_{i = 1}^{n}{\sqrt{2P_{C}}{c_{i}(t)}{\cos\left( {{\omega_{0}t} + \phi_{l}} \right)}}} + {n_{W}(t)}}$where n_(W)(t) is thermal white gaussian noise. The FM receiver filtersthe interference, reducing its power by the processing gain, K, which isdefined as the ratio of the spread-spectrum bandwidth to the FMbandwidth. Typically K=10 MHz/30 kHz=330. The filter output is${v_{f}(t)} = {{\sqrt{2P_{A}}{\cos\left( {{\omega_{0}t} + {\theta(t)}} \right)}} + {\sum\limits_{i = 1}^{n}{\sqrt{2{P_{C}/K}}{z_{i}(t)}{\cos\left( {{\omega_{0}t} + \phi_{l}} \right)}}} + {n(t)}}$where z_(i)(t) is the chip stream c_(i)(t) after filtering, where φ_(l)is the phase of each B-CDMA spread-spectrum interferer relative to theFM carrier, and where the noise n(t) has been filtered.

S. O. Rice, TIME-SERIES ANALYSIS, Chap. 25, John Wiley & Sons, Inc., NewYork, 1963; as well as H. Taub and D. L. Schilling, PRINCIPLES OFCOMMUNICATION SYSTEMS, 2nd Edition, McGraw-Hill, New York, 1986; pointout that the modulation θ(t) has little effect on the calculation of theoutput noise power in an FM output SNR calculation. Setting θ(t) equalto zero for this calculation yields the phasor diagram shown in FIG. 14.

The output of an FM discriminator is the derivative of Ψ with respect totime. Assuming a high input signal to noise ratio, C/I, i.e. assumingthat:P _(A) >>nP _(c) /Kandtan ψ=ψyields$\frac{\mathbb{d}\psi}{\mathbb{d}t} = {\sum\limits_{i = 1}^{n}{\sqrt{\frac{P_{C}/P_{A}}{K}}{z_{i}(t)}\sin\quad{\phi_{l}.}}}$

To simplify the calculations, assume that the heavily filtered chipstream z_(i)(t) approximates bandlimited, white, gaussian noise, with avariance equal to unity. Then, the output interfering power N_(I), dueto the z_(i), in a bandwidth, f_(m)=30 kHz is, according to Taub andSchilling, supra., where B is the intermediate frequency (IF) bandwidth.Similarly, the output power N_(TH) due to thermal noise is:$N_{I} = {{{n\left( \frac{P_{C}/P_{A}}{2K} \right)}\frac{4\pi^{2}}{3B}f_{m}^{3}} = {{n\left( \frac{P_{C}/P_{A}}{2K} \right)}\frac{\omega_{m}^{2}}{3}{f_{m}/B}}}$$N_{TH} = {{\frac{\eta}{P_{A}}\left( \frac{4\pi^{2}f_{m}^{3}}{3} \right)} = {\frac{\eta\quad f_{m}}{P_{A}}{\left( \frac{4\pi^{2}}{3} \right).}}}$

Assuming sine wave modulation, the output signal power is, according toRice, supra.,S _(o)=(βω_(m))²/2

Hence the output signal-to-noise-ratio is $\begin{matrix}{{SNR}_{o} = \frac{S_{o}}{N_{I} + N_{TH}}} \\{= \frac{\beta^{2}{\omega_{m}^{2}/2}}{{{n\left( \frac{P_{C}/P_{A}}{2K} \right)}\omega_{m}^{2}\frac{f_{m}}{3B}} + {\frac{\eta\quad f_{m}}{P_{A}}\left( \frac{\omega_{m}^{2}}{3} \right)}}} \\{= {\frac{3}{2}{{\beta^{2}\left( {{P_{A}/\eta}\quad f_{m}} \right)}/\left\lbrack {1 + {\frac{n}{2K}\left( \frac{P_{C}}{P_{A}} \right)\left( \frac{P_{A}}{{nf}_{m}} \right)}} \right\rbrack}}}\end{matrix}$ or${SNR}_{o} = {\frac{3}{2}{\beta^{2}({CNR})}{\left( \frac{B}{f_{m}} \right)/\left\lbrack {1 + {\frac{n}{2K}\left( \frac{P_{C}}{P_{A}} \right)({CNR})}} \right\rbrack}}$where the average carrier to thermal noise ratio (CNR) equals(P_(A)/ηB). For example, in the AMPS/B-CDMA spread-spectrum overlay ofthe present invention, β=4, B/f_(m)=10, K=330 and P_(A)/P_(C)=10. Thenif, for example, n=144,${SNR}_{o} = {{240{({CNR})/\left\lbrack {1 + {0.22(0.1){CNR}}} \right\rbrack}} = {240{\left( \frac{CNR}{P_{n}} \right)/{\left\lbrack {1 + {0.22({CNR})}} \right\rbrack.}}}}$

FIG. 15 plots SNR_(o) for different numbers of spread-spectrum units, n,in an AMPS antenna sector. FIG. 15 may also be used to estimateperformance in a fading medium. For example, if the number ofsimultaneous B-CDMA spread-spectrum units in an AMPS sector is 144 andif the average carrier to thermal noise ratio, CNR, seen by the AMPSmobile-cellular system is 19 dB, then the demodulated output SNR_(o) ofthe AMPS FM demodulator is 43 dB. A 6 dB fade reduces the CNR to 13 dBand reduce the demodulated output SNR_(o) to 37 dB. The above equationfor SNR_(o) may be used to determine the maximum number of CDMAspread-spectrum units allowable.

Consider a CDMA spread-spectrum communications system receiving adesired spread-spectrum signal in the presence of interference. SinceCDMA spread-spectrum communication systems are limited by interference,rather than limited by thermal noise, the thermal noise may beneglected. As shown in FIG. 6, the spread-spectrum demodulator 161, inthe preferred embodiment, may include as a coherent demodulator 400, asillustrated in FIG. 16, for DS/BPSK CDMA in the presence of CDMA and FMinterference from an AMPS mobile-cellular system. As shown in FIG. 16,the coherent demodulator 400 includes a mixer 402 and an integrator 404.The coherent demodulator 400 is employed to perform the demodulationprocedure. In FIG. 16, the phases θ_(i) and θ_(j) are independent randomvariables uniformly distributed between −π and π, each representing theeffect of random propagation delays. The spreading codes, i.e. chipsequences, are c_(i)(t−τ_(i)) where τ_(i)=0 if the chip edges line up.Assume the chip edges occur displaced from that of unit U_(o) by thetime τ_(i). If one uses a large spreading sequence, each bit appears asthough the bit were spread by a truly random binary sequence and thusτ_(i) can be assumed to be uniformly distributed between 0 and T_(c),where T_(c) is the chip duration.

The ratio T_(b)/T_(c), which is the ratio of a data bit to a chip, iscalled the processing gain. Note that this processing gain is differentfrom the processing gain, K, i.e. the ratio of the spread-spectrumbandwidth to the FM bandwidth. The phase φ_(mj) represents the FMmodulation. In the AMPS mobile cellular system the maximum value of isφ_(mj) approximately equal to 4.

Referring to FIG. 16, one obtains$v_{o} = {{\sqrt{2P_{c}}T_{b}d_{o}} + {\sqrt{2P_{C}}T_{b}{\sum\limits_{i = 1}^{n_{1}}{\cos\quad\theta_{l}\frac{1}{T_{b}}{\int_{0}^{T_{b}}{{c_{o}(t)}{c_{i}\left( {t - \tau_{i}} \right)}{\mathbb{d}t}}}}}} + {\sqrt{2P_{A}}T_{b}{\sum\limits_{j = 1}^{n_{2}}{\frac{1}{T_{b}}{\int_{0}^{T_{b}}{{c_{o}(t)}{\cos\left( {{\phi\quad m\quad{j(t)}} + {\phi\quad j}} \right)}{\mathbb{d}t}}}}}}}$where c_(i)^(′)(t − τ_(i)) = d_(i)(t − τ_(i))c_(i)(t − τ_(i))since the c_(i)'s are assumed random.

For determining the output SNR of the CDMA spread-spectrum communicationsystem, one considers separately the interfering power produced by eachtype of interfering signal.

Interference Caused by Other Spread-Spectrum Units

The interference caused by the ni CDMA spread-spectrum interferers is$I_{1} = {\sqrt{2P_{c}}{\sum\limits_{i = 1}^{n_{1}}{\cos\quad\theta_{l}{\int_{0}^{T_{b}}{{c_{o}(t)}{c_{i}\left( {t - \tau_{i}} \right)}{\mathbb{d}t}}}}}}$The power in I₁, i.e. I₁ ², is$I_{1}^{2} = {\frac{1}{2}{\left( {2P_{c}} \right)\left\lbrack {\sum\limits_{i = 1}^{n_{1}}{\int_{0}^{T_{b}}{{c_{o}(t)}{c_{i}\left( {t - \tau_{i}} \right)}{\mathbb{d}t}}}} \right\rbrack}^{2}}$which uses the fact that$\overset{\_}{\cos^{2}\theta_{l}} = {\frac{1}{2}.}$

M. B. Pursley, “Performance Evaluation For Phase-Coded Spread-SpectrumMultiple-Access Communication—Part I: System Analysis,” IEEETRANSACTIONS ON COMMUNICATIONS, COM-25, No. 8, August, 1977; and othershave shown that the power due to the chips not being in alignment isless than the power obtained when τ_(i)=0, for all i. Indeed theexpression usually used is$\overset{\_}{\left\lbrack {\sum\limits_{i = 1}^{n_{1}}\quad{\int_{0}^{T_{b}}{{c_{o}(t)}{c_{i}^{\prime}\left( {t - \tau_{i}} \right)}\quad{\mathbb{d}t}}}} \right\rbrack^{2}} = {\frac{2}{3}{n_{1}.}}$

The value (⅔) n₁ obtained above is dependent on the pulse shape andvaries for the cases considered: ⅔ (rectangular), ¾ (half-sine wave) and0.8 (raised-cosine). While the derivation of this result is, in general,complicated, the following three simple examples illustrate of thederivation.

Rectangular Pulses

Consider the rectangular pulses c.sub.o and c′.sub.i illustrated in FIG.17. Note that the product c_(o)(t) c′_(i)(t−τ_(i)) is${{c_{o}(t)}{c_{i}^{\prime}\left( {t - \tau_{i}} \right)}} = \left\{ {{\begin{matrix}{\pm 1} & {{0 \leq t \leq \tau_{i}},{{{each}\quad{with}\quad P} - \frac{1}{2}}} \\{\pm 1} & {{\tau_{i} \leq t < T_{c}},{{{each}\quad{with}\quad P} = {\frac{1}{2}.}}}\end{matrix}{Therefore}{\int_{0}^{T_{c}}{{c_{o}(t)}{c_{i}^{\prime}\left( {t - \tau_{i}} \right)}\quad{\mathbb{d}t}}}} = \left\{ {{\begin{matrix}{{{\pm T_{c}}\quad{each}\quad{with}\quad P} = \frac{1}{2}} \\{{{\pm \left( {T_{c} - \tau_{i}} \right)}\quad{each}\quad{with}\quad P} = \frac{1}{2}}\end{matrix}{and}\quad{so}\overset{\_}{{{\int_{0}^{T_{c}}{{c_{o}(t)}{c_{i}^{\prime}\left( {t - \tau_{i}} \right)}\quad{\mathbb{d}t}}}}^{2}}} = \overset{\_}{{{\frac{1}{2}\left( T_{c} \right)^{2}} + {\frac{1}{2}\left( {T_{c} - \tau_{i}} \right)^{2}}} = {\frac{2}{3}T_{c}^{2}}}} \right.} \right.$if one assumes that τ_(i) was uniformly distributed between 0 and T_(c).Note that if τ_(i)=0,${\int_{0}^{T_{c}}{{c_{o}(t)}{c_{i}^{\prime}(t)}\quad{\mathbb{d}t}}} = {{{\pm T_{c}}\quad{each}\quad{with}\quad P} = \frac{1}{2}}$and$\overset{\_}{\left\lbrack {\int_{0}^{T_{c}}{{c_{0}(t)}{c_{i}^{\prime}(t)}\quad{\mathbb{d}t}}} \right\rbrack^{2}} = {T_{c}^{2}.}$

Ergo, comparing the above equations yields a factor of ⅔.

Half-Sinusoidal Input Pulses

A second example assumes c′_(i)(t)=sin [πt/(Tc)], and, for simplicity ofcalculation, c_(o)(t) is taken to be rectangular as illustrated in FIG.18. In this example,${\int_{0}^{T_{c}}{{c_{o}(t)}{c_{i}\left( {t - \tau_{i}} \right)}\quad{\mathbb{d}t}}} = {{\pm {\frac{T_{c}}{\pi}\left\lbrack {1 - {\cos\quad\frac{{\pi\pi}_{i}}{T_{c}}}} \right\rbrack}} \pm {{\frac{T_{c}}{\pi}\left\lbrack {{\cos\quad\frac{{\pi\pi}_{i}}{T_{c}}} - 1} \right\rbrack}.}}$

The average power is then $\begin{matrix}{\overset{\_}{\left\lbrack {\int_{0}^{T_{c}}{{c_{o}(t)}{c_{i}\left( {t - \tau_{i}} \right)}\quad{\mathbb{d}t}}} \right\rbrack^{2}} = {\frac{1}{T_{c}}{\int_{0}^{T_{c}}{\begin{bmatrix}{{\frac{1}{2}\left( \frac{2T_{c}}{\pi} \right)^{2}} + {\frac{1}{2}\left( \frac{2T_{c}}{\pi} \right)^{2}}} \\{\cos^{2}\frac{{\pi\pi}_{i}}{T_{c}}}\end{bmatrix}\quad{\mathbb{d}\tau_{i}}}}}} \\{= {\frac{3}{4}{\left( \frac{2T_{c}}{\pi} \right)^{2}.}}}\end{matrix}$

If τ_(i)=0, then it is readily shown that$\overset{\_}{\left\lbrack {\int_{0}^{T_{c}}{{c_{0}(t)}{c_{i}^{\prime}(t)}\quad{\mathbb{d}t}}} \right\rbrack^{2}} = {\left( \frac{2T_{c}}{\pi} \right)^{2}.}$

Note the factor now becomes ¾ rather than ⅔.

The Raised-Cosine

If c′_(i)(t)=1−cos [2πt/(T_(c))], i.e. a raised cosine, 0.8T_(c) ²results compared to T_(c) ² if τ_(i)=0, and the factor is now 0.8. Forrectangular shape pulses, the equation for the power in I_(i) becomes$\overset{\_}{I_{i}^{2}} = {\frac{T_{b}^{2}n_{1}}{3}\left( {2P_{c}} \right)\frac{f_{b}}{f_{c}}}$where the denominator of 3 is derived by well known techniques in theprior art, as well as by alternative derivations of the error rate inCDMA spread-spectrum system; for example, by J. M. Holtzman, “A Simple,Accurate Method To Calculate Spread-Spectrum Multiple-Access ErrorProbabilities,” IEEE TRANSACTIONS ON COMMUNICATIONS, vol. 40, No. 3,March 1992.

Interference Caused by the FM Units

The interference caused by the n2 FM AMPS mobile-cellular units is$I_{2} = {\sqrt{2P_{A}}T_{b}{\sum\limits_{j = 1}^{n_{2}}\quad{\frac{1}{T_{b}}{\int_{0}^{T_{b}}{{c_{o}(t)}\quad\cos\quad\left( {{\phi_{mj}(t)} + \phi_{j}} \right)\quad{{\mathbb{d}t}.}}}}}}$

To simplify the discussion one approximates I₂ by the summation$I_{2} = {\sqrt{2P_{A}}T_{c}{\sum\limits_{j = 1}^{n_{2}}\quad{\sum\limits_{i = 1}^{T_{b}/T_{c}}\quad{{C_{o}\left( {iT}_{c} \right)}\quad{{\cos\quad\left\lbrack {{\phi_{mj}\left( {iT}_{c} \right)} + \phi_{j}} \right\rbrack}.}}}}}$

Then the variance of I₂ becomes$\overset{\_}{I_{2}^{2}} = {\left( {2P_{A}} \right)T_{c}^{2}{\sum\limits_{j = 1}^{n_{2}}\quad{\sum\limits_{i = 1}^{T_{b}/T_{c}}\quad{E{\left\{ {\cos^{2}\left\lbrack {{\phi_{mj}\left( {iT}_{c} \right)} + \phi_{j}} \right\rbrack} \right\}.{Hence}}}}}}$$\overset{\_}{I_{2}^{2}} = {T_{b}^{2}{n_{2}\left( P_{A} \right)}{f_{b}/{f_{c}.}}}$

Signal to-Noise Ratio

From the equations for v_(o) and the power in I₁ and I₂, the SNR seen bya B-CDMA spread-spectrum unit in the presence of other B-CDMAspread-spectrum units and AMPS units becomes: $\begin{matrix}{{SNR}\quad = \quad\frac{2\quad P_{\quad C}\quad T_{\quad b}^{\quad 2}}{\quad{{2\quad P_{\quad C}\quad T_{\quad b}^{\quad 2}\quad\left( \quad{\frac{\quad n_{\quad 1}}{\quad 3} \cdot \quad\frac{\quad f_{\quad b}}{\quad f_{\quad c}}} \right)}\quad + \quad{2\quad P_{\quad A}\quad T_{\quad b}^{\quad 2}\quad\left( \quad{\frac{\quad n_{\quad 2}}{\quad 2} \cdot \quad\frac{\quad f_{\quad b}}{\quad f_{\quad c}}} \right)}}}} \\{\quad{= \quad{\frac{\quad{f_{\quad c}/\quad f_{\quad b}}}{\quad{\frac{\quad n_{\quad 1}}{\quad 3}\quad + \quad{\frac{\quad n_{\quad 2}}{\quad 2}\quad\left( \quad\frac{\quad P_{\quad A}}{\quad P_{\quad C}} \right)}}}\quad.}}\quad}\end{matrix}$

The Cellular Overlay

Using the equations for the power in I₂, the operation of the cellularoverlay is indicated by the effect of AMPS units on the spread-spectrumbase station, as shown in FIG. 19. Assuming that the spread-spectrumbase station employs sector means with a 60 degree, or six sector,antenna, the number of interfering AMPS units isn ₂=1.6(50/6)=13.where 50 is the number of AMPS mobile cellular units in a cell, six isthe fraction of units in the sector and the factor 1.6 includes adjacentcell interference.

The processing gain f_(c)/f_(b)=10 Mchips/13 (kbits/sec) or$\frac{f_{c}}{f_{b}} = {\frac{10\quad{{Mchips}/s}}{13\quad{{kbits}/s}} = 770.}$

Let the ratio of the powers transmitted by the mobile-cellular units andB-CDMA spread-spectrum units be P_(A)/P_(C)=15, where P_(A)/P_(C) is theratio of the power received at a base station due to an AMPS unit to thepower received at the base station due to a CDMA spread-spectrum unit.Then the power in I₂ becomes:${SNR} = {\frac{770}{\frac{n_{1}}{3} + {\frac{13}{2}(15)}}.}$

If the SNR at the receiver is 4 dB, or 6 dB, which is sufficient toyield an error rate of 10⁻³, since the adaptive delta modulator (ADM)performs properly with a BER=10⁻², then n₁=285, neglecting VAD.

If approximately 60% of the interfering units are in the adjacentsectors of the neighboring two cells and since there are six sectors,more than (285/1.6)×6=1069 CDMA spread-spectrum units can simultaneouslyaccess the CDMA spread-spectrum base station neglecting VAD and 2137CDMA spread-spectrum units with VAD.

Unit interference to the base station does not limit performance by theeffect of B-CDMA spread-spectrum units on the AMPS mobile-cellular basestation, as shown in FIG. 20. Assuming that the AMPS mobile-cellularbase station uses a three-sector antenna, the equation for SNR_(o)yields the FM demodulated output SNR:$({SNR})_{o} = {\frac{3}{2}{\beta^{2}({CNR})}{\left( \frac{B}{f_{m}} \right)/{\left\{ {1 + {\frac{n_{1}}{2K}\left( \frac{P_{C}}{P_{A}} \right)({CNR})}} \right\}.}}}$where CNR is the carrier power-to-thermal noise ratio.

Let β=4, B/f_(m)=10, K=330, P_(A)/P_(C)=10 and CNR=50=17 dB. If thesystem is designed so that the CDMA interference is equal to the thermalnoise power, then or 198 units neglecting VAD.n ₁=2K(P _(A) /P _(C))/CNR=2(330)(15)/50=198

Again, assuming that 60% of the interference comes from units inadjacent cells and including VAD, the number of B-CDMA spread-spectrumunits in the cell, from all three sectors, is$N_{CDMA} = {{\frac{198}{1.6} \times {\overset{2}{\underset{VAD}{\uparrow}}{\times \overset{3}{\underset{Sectors}{\uparrow}}}}} = 742.}$

Other values for different C/N ratios are readily obtained from FIG. 15.

Also, the effect of the B-CDMA spread-spectrum-base station and the AMPSmobile-cellular base stations on CDMA spread-spectrum units, as shown inFIGS. 21 and 22, respectively. The SNR seen by a CDMA spread-spectrumunit is given by the equation for the power in I₂. However, in the worstcase, a CDMA spread-spectrum unit is in the corner of three cells andthe power reaching the worst case CDMA spread-spectrum unit from the3-AMPS base mobile-cellular stations is n₂ P_(A) where$n_{2} = {{\underset{3\quad{sectors}}{\overset{({50/3})}{\uparrow}}\underset{3\quad{base}\quad{stations}}{\overset{(3)}{\uparrow}}} = 50.}$

The power received by the worst-case CDMA spread-spectrum unit from eachof the three spread-spectrum base stations is n₁{overscore (P)}, where{overscore (P)} is the average power received by a B-CDMAspread-spectrum unit. Each CDMA spread-spectrum base station transmits apower of P₁=0.77 {overscore (P)} Watts/unit to two thirds of itsspread-spectrum units near the spread-spectrums base station and thepower 1.44 {overscore (P)} to the one-third of its units that are farfrom the spread-spectrum base station. This constitutes a coarse forwardpower control. Let us also set P_(A)/{overscore (P)}=9.4 in the forwarddirection. Using the equation for the power in I₂ for this worst casecondition yields:${SNR} = {\frac{1.44 \times 770}{\frac{n_{1}}{3} + {\frac{50}{2}(9.4)}}.}$

For a SNR of 6 dB, or BER=10⁻³, one determines n₁=127 units neglectingVAD. By including VAD, by including the fact that the interference iscaused by three cells, and by including there are six sectors in a cell,the maximum number of spread-spectrum units is:N ₁=6(sectors)·2(VAD)n ₁/3(cells)=4n ₁=506.

Referring to FIGS. 21 and 22, it is readily shown that for the units inthe ⅔ area closest to the base station, the SNR is${SNR} = {\frac{0.77 \times 770}{\frac{n_{1}}{3} + {\frac{50 \times {1.6/3}}{2}(9.4)}}.}$

Note the 50×1.6/3 in the denominator of the above equation representsthat the AMPS mobile-cellular base station in the cell is contributinginterference to spread-spectrum units near the spread-spectrum basestation and 60% additional interference is coming from base stations inneighboring cells. Solving the equation for the SNR yields n₁=71.7.

Since the B-CDMA interference comes from the CDMA spread-spectrum basestation in the cell and also an additional 60% comes from adjacent basestations,N ₁=6(sectors)·2(VAD)n ₁/1.6=7.5 n ₁=538where the factor 1.6 includes adjacent cell interference.

Also, the effect of the CDMA spread-spectrum base stations on AMPSmobile-cellular units is shown in FIG. 23. The SNR seen by the AMPSmobile-cellular units is obtained from the equation from the equationfor SNR_(o) above, resulting in:${SNR}_{0} = {\frac{3}{2}{\beta^{2}({CNR})}{\left( \frac{B}{f_{m}} \right)/{\left( {1 + {\frac{n_{1}}{2K}\left( \frac{\overset{\_}{P}}{P_{A}} \right)({CNR})}} \right).}}}$

Then, with P_(A)/{overscore (P)}=9.4 and K=330 and CNR=50=17 dB, thevalue of N₁ such that the CDMA interference is equal to the AMPSinterference is$n_{1} = {{2{K/\left( \frac{\overset{\_}{P}}{P_{A}} \right)}({CNR})} = {\frac{660(9.4)}{(50)} = 124}}$or 124 units in the three sectors without VAD.

The maximum number of CDMA spread-spectrum units in the cell is then$N_{1} = {{{124 \cdot \frac{1}{3} \cdot 6}{({sectors}) \cdot 2}({VAD})} = 496}$or 496 units, where the factor of ⅓ takes into account three basestations.

The results of these calculations indicate that the limitation on thenumber of units is the interference produced by the B-CDMAspread-spectrum base stations to a worst-case AMPS mobile-cellular unitlocated in the corner of three cells. Even with this limitation, 496B-CDMA spread-spectrum units can overlay on top of 50 AMPSmobile-cellular units. Thus the combined total number of units in the 10MHz band become 496+50=578 units. This result is approximately eleventimes the original number of AMPS mobile-cellular units using thespectrum.

Note that exactly the same calculations hold for TDMA. In the TDMA case,however, the total number of units is 496+150=646 which is a factor 4.3times greater then the original number of TDMA units.

Consider the effect of the spread-spectrum-base station on a cellularunit. The power of the spread-spectrum signal from thespread-spectrum-base station is spread over 10.0 MHz. The cellular unit,however, communicates on a predetermined channel using FM, which has abandwidth of approximately 30 kHz. Thus, the cellular unit has aneffective processing gain with respect to the spread-spectrum signalfrom the spread-spectrum-base station of approximately 330, or 25 dB.The 25 dB means that the power level of the spread-spectrum signal fromthe spread-spectrum-base station is reduced at the cellular unit by 330.Assuming that the spread-spectrum-base station and cellular-base stationeach have a transmitter power level of 10 Watts, the processing gainyields an acceptable signal-to-interference ratio at the cellular unit,i.e. much higher then the required 17 dB.

The effect of the cellular-base station from the spread-spectrum-basestation as follows: The spread-spectrum signal from thespread-spectrum-base station is spread by the chip rate of 6.25megachips per second. The data rate of the data in the spread-spectrumsignal is 4,800 bits per second. Thus, the processing gain at thespread-spectrum unit is 6.25 megachips per second divided by 4,800 bitsper second, which approximately equals 1,250, or approximately 31 dB.Assuming the spread-spectrum-base station and the cellular-base stationeach have a transmitter power of 10 Watts, this processing gain yieldsan acceptable signal-to-interference ratio at the spread-spectrum unit,i.e. 31 dB.

Consider the effect of spread-spectrum units on the receiver at thecellular-base station. Assume, for ease of calculations, that units ofthe mobile-cellular system and units of the spread-spectrum systememploy adaptive power control. The cellular unit transmits a power,P_(CELL)=0.5 W, and the spread-spectrum unit transmits a power P_(SS)=10mW. Each cell of a mobile-cellular system is assumed to have 50 cellularunits, and the spread-spectrum system is assumed to have K units. Theinterference to the receiver of the cellular-base station is N timesP_(SS) divided by the processing gain. As shown before the processinggain is N=10.0 MHz/30 kHz=330 or 25 dB. Thus, the signal-to-interferenceratio isNP _(CELL)/(K×P _(SS))=330(½)/K(2.01)=2×10₄ /K.Assuming 200 spread-spectrum units (K=200), yields asignal-to-interference ratio of 20 dB, which exceeds the 17 dB signal tointerfere ratio required for the FDMA used today and greatly exceeds the7 dB signal-to-interference ratio needed in the projected TDMA system.The presently deployed mobile-cellular system typically has P_(CELL)=500mW for hand held telephones and P_(CELL) equals one Watt for automobilelocated telephones. Thus, the foregoing analysis requires that thespread-spectrum unit transmits a power level of ten mW, P_(SS)=10 mW.

Consider the effect of the foregoing power levels on thespread-spectrum-base station. The spread-spectrum-base station receivesan interfering power level from 50 cellular units, of 50 times one Watt.With a processing gain for the spread-spectrum system of N=1250, asignal-to-interference ratio results at the spread-spectrum-base stationof S/I=(10 mW×1250)/(1 W×50), yielding S/I=¼ which is −6 dB. Thereceiver at the spread-spectrum-base station requires a signal to noiseratio of 4 dB. The required SNR can be realized at thespread-spectrum-base station with a band reject filter for notching outthe signals from the cellular units in the 30 kHz predeterminedchannels. With a properly designed comb-notch filter, a 20 dB to 30 dBsignal-to-interference ratio can readily be achieved.

FIG. 24 illustrates a comb-notch filter 333 inserted in a receiver of aspread-spectrum-base station. The receiver includes a low noiseamplifier 331 coupled between the antenna 330 and a down converter 332.The comb-notch filter 333 is coupled between the down converter 332 andbase-spread-spectrum demodulator 334. A base-demodulator 335 is coupledto the base-spread-spectrum detector 334. In the preferred embodiment,the base-demodulator 334 may include a coherent demodulator 400, asillustrated in FIG. 16, for DS/BPSK CDMA in the presence of CDMA and FMinterference from an AMPS system. As shown in FIG. 16, the coherentdemodulator 400 includes a mixer 402 and an integrator 404. The coherentdemodulator 400 is employed to perform the demodulation procedure. Thecomb-notch filter 333 in this illustrative example operates at anintermediate frequency and removes interference from the mobile-cellularsystem.

In the preferred embodiment, the present invention generates a 10 MHzwide spread-spectrum signal, and employs a six segment antenna, as shownin FIG. 25. In each segment, the power transmitted at the frequenciesused for transmission by the three intersecting cells will be notchedout, as shown in FIG. 2. As illustratively shown in FIG. 26, the powertransmitted in Cell 1, Cell 2, and Cell 3 is removed from the shadedregion. In addition, the spread-spectrum-base station receiver notchesout each 30 kHz channel used by the cellular users in Cell 1, forexample, as illustrated in FIG. 2. Since, in the shaded region of FIG.26, the B-CDMA spread-spectrum base stations in cell one, cell two, andcell three do not transmit in the frequency bands used by the cellularuser shown, for example, in the shaded region of FIG. 26, the cellularuser receives no interference.

From the foregoing analysis, a person of skill in the art recognizesthat the present invention allows a spread-spectrum CDMA system tooverlay on a pre-existing FDMA mobile-cellular system, withoutmodification to the pre-existing mobile-cellular system. The presentinvention allows frequency reuse of the already allocated frequencyspectrum to the mobile-cellular system. At the same time performance ofthe mobile-cellular system is not degraded. The spread-spectrum systemmay add an increase of 200 spread-spectrum units over the 50 cellularunits. The present system performance calculations are consideredconservative, and an increase in spread-spectrum units may be greaterthan the estimated 200.

It will be apparent to those skilled in the art that variousmodifications can be made to the spread-spectrum CDMA communicationssystem of the instant invention without departing from the scope orspirit of the invention, and it is intended that the present inventioncover modifications and variations of the spread-spectrum CDMAcommunications system provided they come in the scope of the appendedclaims and their equivalents.

1. A receiving apparatus comprising: an antenna receiving at least onecellular code division multiple access (CDMA) signal, each receivedcellular CDMA signal having an associated spread bandwidth; and a notchfilter device configured to attenuate a bandwidth at a plurality offrequencies within the associated spread bandwidth of the at least onereceived cellular CDMA signal.
 2. The receiving apparatus of claim 1wherein the attenuated bandwidth at each of the plurality of frequenciesdoes not exceed 30 kHz.
 3. The receiving apparatus of claim 1 whereinthe attenuated bandwidth at each of the plurality of frequencies is notless than 6 kHz.
 4. The receiving apparatus of claim 1 wherein theattenuated bandwidth at each of the plurality of frequencies is in arange from 6 kHz to 30 kHz.
 5. The receiving apparatus of claim 4wherein the associated spread bandwidth does not exceed 10 MHz.
 6. Thereceiving apparatus of claim 1 wherein the notch filtering device is acomponent of a mobile unit cellular CDMA receiver system.
 7. Thereceiving apparatus of claim 6 wherein the cellular CDMA receiver systemis a component of a cellular base station.
 8. The receiving apparatus ofclaim 1 wherein the receiving apparatus is configured to receive signalsin a time division multiple access (TDMA) format.
 9. The receivingapparatus of claim 1 further comprising a down converter for convertingthe at least one cellular CDMA signal to an intermediate frequency (IF)signal and providing the IF signal to the notch filter.
 10. Thereceiving apparatus of claim 1 further comprising a despreader fordespreading an output of the notch filter device.
 11. A methodcomprising: receiving at least one cellular code division multipleaccess (CDMA) signal, each received cellular CDMA signal having anassociated spread bandwidth; and attenuating a bandwidth at a pluralityof frequencies within the associated spread bandwidth of the at leastone received cellular CDMA signal.
 12. The method of claim 11 whereinthe attenuated bandwidth at each of the plurality of frequencies doesnot exceed 30 kHz.
 13. The method of claim 11 wherein the attenuatedbandwidth at each of the plurality of frequencies is not less than 6kHz.
 14. The method of claim 11 wherein the attenuated bandwidth at eachof the plurality of frequencies is in a range from 6 kHz to 30 kHz. 15.The method of claim 14 wherein the associated spread bandwidth does notexceed 10 MHz.
 16. The method of claim 11 where the method is performedby a component of a cellular CDMA receiver system.
 17. The method ofclaim 16 wherein the method is performed by a cellular CDMA receiversystem which is a component of a cellular base station.
 18. The methodof claim 11 further comprising receiving signals in a time divisionmultiple access (TDMA) format.
 19. The method of claim 11 furthercomprising converting the at least one cellular CDMA signal to anintermediate frequency (IF) signal and the attenuating is performed onthe IF signal.
 20. The method of claim 11 further comprising despreadinga result of the attenuating.
 21. A notch filtering apparatus comprising:an input configured to receive at least one cellular code divisionmultiple access (CDMA) signal, each received cellular CDMA signal havingan associated spread bandwidth; and a notch filter device configured toattenuate a bandwidth at a plurality of frequencies within theassociated spread bandwidth of the at least one received cellular CDMAsignal.
 22. The notch filtering apparatus of claim 21 wherein theattenuated bandwidth at each of the plurality of frequencies does notexceed 30 kHz.
 23. The notch filtering apparatus of claim 21 wherein theattenuated bandwidth at each of the plurality of frequencies is not lessthan 6 kHz.
 24. The notch filtering apparatus of claim 21 wherein theattenuated bandwidth at each of the plurality of frequencies is in arange from 6 kHz to 30 kHz.
 25. The notch filtering apparatus of claim24 wherein the associated spread bandwidth does not exceed 10 MHz. 26.The notch filtering apparatus of claim 21 wherein the notch filteringdevice is a component of a cellular CDMA receiver system.
 27. The notchfiltering apparatus of claim 26 wherein the cellular CDMA receiversystem is a component of a cellular base station.